Generating two signals having a mutual phase difference of 90°

ABSTRACT

A method and device are described for generating two output signals (I; Q) each substantially identical to a square-wave input signal (A) from a local oscillator ( 2 ), wherein the first output signal (I) may have a certain time shift with respect to the input signal (A), and wherein the second output signal (Q) is shifted over T 1 /4 [mod T 1 ] with respect to the first output signal (I), T 1  being the period of the input signal (A). To generate the first output signal (I), Fourier components (S 1 (ω 1 ), S 3 (ω 3 ), S 5 (ω 5 ), S 7 (ω 7 ), S 9 (ω 9 ), S 11 (ω 11 ) etc) of the input signal are combined. To generate the second output signal (Q), Fourier components (S 1 (ω 1 ), S 5 (ω 5 ), S 9 (ω 9 ) etc) of the input signal are phase shifted over +90° while Fourier components (S 3 (ω 3 ), S 7 (ω 7 ), S 11 (ω 11 ) etc) of the input signal are phase shifted over −90°, and the thus shifted Fourier components of the input signal are combined.

[0001] In many fields, there is a general desire to be able to generatetwo signals with a mutual 90° phase difference, both substantiallyidentical to one input signal. Examples of fields where such signalcombination is desirable are for instance: television tuners, mobiletelephones (GSM, NMT), wireless telephones (DECT), etc. As is commonlyknown, quadrature signals are used in, for instance, television tunersfor, inter alia, mirror rejection.

[0002] For generating such signal combination as mentioned above,several techniques are already available. Each of those availabletechniques suffers from some disadvantages. One example of such knowntechnique is to use frequency division with gate circuits, flip flops,etc. A disadvantage of this known technique is that the frequency of theinput signal must be chosen at least twice as high as the desiredfrequency of the 0° and 90° signals. Another technique is to use aseparate oscillator which generates two output signals having 90° phasedifference, the oscillator being coupled to the original input signal bymeans of a phase locked loop. A disadvantage of this technique is thatit involves a relatively large amount of electronic circuitry and arelatively large amount of energy dissipation.

[0003] A further technique that is known per se is to use aphase-shifting network. Phase-shifting networks generate one outputsignal identical to the input signal, with a fixed phase delay that canbe set to 90°. However, a disadvantage of phase-shifting networks todate is that they require the input signal to be sine-shaped. Moreparticularly, phase-shifting networks to date are not capable ofgenerating two output signals having a mutual 90° phase difference andboth at least substantially identical to an input signal having a squarewave shape, although most local oscillators in the above-mentionedexamples generate a square wave signal.

[0004] The present invention aims to provide an approach which allowsthe use of a phase-shifting network to generate two output signalshaving a mutual 90° phase difference, without the restriction that theinput signal needs to be sine-shaped. To this end, the inventionprovides a method and device for generating two signals having a mutualphase difference of 90° as defined in the independent claims. Thedependent claims define advantageous embodiments. In a preferredembodiment, the present invention provides a phase-shifting networkcapable of receiving a square wave signal and of outputting two squarewave signals having a mutual 90° phase difference and both at leastsubstantially identical to the input signal.

[0005] The invention is based on the insight that a square wave signalsuch as generated by a local oscillator can be approximated as a Fourierseries of a fundamental wave and a limited number of odd harmonic waves,each of said waves being sine-shaped. In practice, the number of oddharmonic waves that need to be taken into account depends on thefrequency of the fundamental wave: the higher the frequency of thefundamental wave, the lower the number of odd harmonic waves that play asignificant role.

[0006] Therefore, based on this insight, the present invention proposesa phase-shifting network which operates on the fundamental Fouriercomponent and at least the third harmonic Fourier component, andpreferably also the fifth harmonic Fourier component, of an input signalin such a way that these components are shifted over the same amount oftime equal to one fourth of the period of the fundamental Fouriercomponent.

[0007] In a preferred embodiment, the phase-shifting network of theinvention comprises a passive polyphase filter. Such filter has anintrinsic property of shifting all frequency components within a passband over the same angle. Assuming that the desired shift angle is equalto 90°, said intrinsic property would be correct for the fifth harmonicFourier component but would be wrong for the third harmonic Fouriercomponent, which should be shifted over 270°, which is equivalent to ashift over −90°. The invention is based on the further insight that thisis equivalent to a +90° shift of the third harmonic Fourier componentwith negative frequency. Therefore, according to the invention in thispreferred embodiment, the phase-shifting network has a frequencycharacteristic that passes the third harmonic Fourier component withnegative frequency and suppresses the third harmonic Fourier componentwith positive frequency.

[0008] These and other aspects, characteristics and advantages of thepresent invention will be further clarified by the following descriptionof exemplary embodiments of a network in accordance with the invention.

[0009] In the drawings, in which same reference numerals indicate equalor similar parts,

[0010]FIG. 1 schematically shows a polyphase filter;

[0011] FIGS. 2A-C illustrate how a polyphase filter can be used togenerate a twophase output signal with positive frequency on the basisof a one-phase input signal;

[0012] FIGS. 3A-C illustrate the shifting of Fourier components;

[0013]FIG. 4 is a block diagram schematically illustrating an embodimentof a device according to the invention;

[0014] FIGS. 5A-C illustrate how a polyphase filter can be used togenerate a twophase output signal with negative frequency on the basisof a one-phase input signal;

[0015]FIG. 6 shows the frequency characteristic of a polyphase filteraccording to the present invention; and

[0016]FIG. 7 schematically illustrates a simple embodiment of apolyphase filter according to the present invention.

[0017]FIG. 1 schematically shows a polyphase filter IO having a firstinput 1l, a second input 12, a first output 13 and a second output 14.More particularly, this polyphase filter 10 is a two-phase filter.Polyphase filters are known per se. For instance, reference is made tothe article “A Fully Integrated 900 MHz CMOS Double QuadratureDownconvertor” by J. Crols et al. in 1995 ISSCC Digest of TechnicalPapers, Vol.38, IEEE press, 1995, p. 136-137. Therefore, an elaborateexplanation of the design and operation of the polyphase filter 10 isnot necessary here. However, in order to introduce some symbols andexpressions, some aspects of the operation of the polyphase filter 10(two-phase filter) are described here.

[0018] Assume that two input signals X₁₁(ω) and X₁₂(ω) are applied tothe two inputs 11 and 12, respectively, the two input signals X₁₁(ω) andX₁₂(ω) being sine-shaped and having the same frequency ω, but having aphase difference of 90°. This can be written as |φ₁₁−φ₁₂|=90° [mod360°], wherein φ₁₁ is the phase of the first input signal X₁₁(ω) appliedto the first input 11, while φ₁₂ is the phase of the second input signalX₁₂(ω) applied to the second input 12. Two situations can bedistinguished:

[0019] 1) X₁₁(ω) is leading, i.e. φ₁₁−φ₁₂=+90°

[0020] 2) X₁₂(ω) is leading, i.e. φ₁₁−φ₁₂=−90°

[0021] A sine-shaped signal may be represented in complex notation asX(ω)=X·e^(jωt), keeping in mind that the actual physical signal is thereal part of the complex expression. Then, both the above-mentionedcases of φ₁₁−φ₁₂=+90° and φ₁₁−φ₁₂=−90° can be written as

X ₁₁(ω)=X·e ^(jωt) and X ₁₂(ω)=jX·e ^(jωt)

[0022] Using X=|X|·e^(jφ) yields:

Re(X ₁₁)=|X|cos(ωt+φ) and Re(X ₁₂)=|X|cos(ωt+φ+π/2)

[0023] so that ω<0 corresponds to the case of φ₁₁−φ₁₂=+90°, whereas ω>0corresponds to the case of φ₁₁−φ₁₂=−90°.

[0024] At its outputs 13 and 14, the polyphase filter 10 generatessine-shaped output signals Y₁₃(ω) and Y₁₄(ω), respectively, having thesame frequency ω as the two input signals X₁₁(ω) and X₁₂(ω).

[0025] It is further assumed that the polyphase filter 10 has a transfercharacteristic H(ω) that can be described as

H(ω)=Y ₁₃ /X ₁₁ =Y ₁₄ /X ₁₂  (1)

[0026] in the case that X₁₂(ω)=jX₁₁(ω).

[0027] If the (normalized) transfer characteristic H(ω) of the polyphasefilter 10 is such that, for a certain positive frequency ω_(x), thefollowing equation (2) is valid:

H(ω_(x))=1 and H(−ω_(x))=0  (2)

[0028] then the polyphase filter 10 can be used for generating twooutput signals Y₁₃ and Y₁₄=jY₁₃ on the basis of only one input signalX₁₁, as will be explained with reference to FIGS. 2A-C.

[0029] In FIG. 2A, a first input signal X_(11A)=X(ω_(x)) is applied tothe first input 11, and a second input signal X_(12A)=iX_(11A)=jX(ω_(x))is applied to the second input 12. It follows from the above equations(1) and (2) that the polyphase filter 10 then generates a first outputsignal Y_(13A)=Y(ω_(x)) and a second input signalY_(14A)=jY_(13A)=jY(ω_(x)) at its two outputs 13 and 14, respectively.

[0030] In FIG. 2B, a first input signal X_(11B)=X(ω_(x)) is applied tothe first input 11 and a second input signalX_(12B)=−jX_(11B)=−jX(ω_(x)) is applied to the second input 12. Itfollows from the above equations (1) and (2) that the polyphase filter10 then generates zero output signals Y_(13B)=0 and Y_(14B)=0 at its twooutputs 13 and 14, respectively.

[0031] The polyphase filter 10 is a linear filter, which means that, iftwo input signals are added, the corresponding output signals are alsoadded. In FIG. 2C, the two input signals used in FIG. 2A and in FIG. 2B,respectively, are added. In other words, the first input 11 receives

X _(11C) =X _(11A) +X _(11B)=2X(ω_(x)),

[0032] while the second input 12 (not connected) receives

X _(12C) =X _(12A) +X _(12B) =jX(ω_(x))+(−jX(ω_(x)))=0.

[0033] Then, also the two output signals of FIG. 2A and FIG. 2B areadded; therefore, the polyphase filter 10 then generates a first outputsignal Y_(13C)=Y(ω_(x)) and a second output signal Y_(14C)=jY(ω_(x)) atits two outputs 13 and 14, respectively.

[0034] In other words, if the first input 11 receives a real inputsignal X(ω_(x)) while the second input 12 is zero, the polyphase filter10 then generates first and second output signals Y₁₃(ω_(x))=½·X(ω_(x))and Y₁₄(ω_(x))=½−jX(ω_(x)). These are two real output signals with 90°phase difference, also indicated as a sine-shaped two-phase outputsignal having positive frequency. In the above, any possible phasedifference between Y₁₃(ω_(x)) and X(ω_(x)) is neglected.

[0035] It is noted that the above explanation would also apply in casethe sign of Y_(14A) would be reversed. In that case, the two-phaseoutput signal may be indicated as having negative frequency. However, itis also possible to consider output 14 as the “first” output and toconsider output 13 as the “second” output.

[0036] Thus, using a polyphase filter, it is possible to generate twosignals Y₁₃(ω_(x)) and Y₁₄(ω_(x)) with 90° phase difference with respectto each other, on the basis of one input signal X(ω_(x)).

[0037] In the above, two assumptions have been made. One assumption isthat the input signal X(ω_(x)) is sine-shaped. The second assumption isthat the frequency ω_(x) of the input signal X(ω_(x)) lies in afrequency region where equation (2) is valid. Such frequency region willhereinafter also be indicated as opposite sign rejecting pass region,abbreviated as OSR pass region, indicating that frequencies within theregion are passed whereas identical frequencies with opposite sign arerejected or at least suppressed.

[0038] To date, broadband polyphase filters exist where the OSR passregion ranges from 0 to very high frequencies, approximating infinity,at least for all practical purposes. Also, polyphase filters have beendesigned for cooperation with a specific local oscillator operating in aspecific frequency band with central frequency ω_(LO) and bandwidthBW_(LO); those polyphase filters have an OSR band-pass region coincidingwith the operational frequency band of the local oscillator, thetransfer function H(ω) being zero for all other frequencies.

[0039] With such prior art polyphase filters it is, however, notpossible to employ the same simple approach in generating two signalswith 90° phase difference with respect to each other, on the basis ofone input signal, if the input signal is a binary signal and the outputsignals are required to be binary signals, too. More specifically, inmany applications, the local oscillator generates a square wave signalwith 50% duty cycle; the technique described in the above can not beused in such cases. This will be explained in the following.

[0040] Assume that the local oscillator generates an output signal Abeing a square wave signal with 50% duty cycle, having a period T₁, asillustrated in FIG. 3A. As is well known, such square wave signal A canbe developed into sine-shaped signal components (Fourier series). Thesesine-shaped signal components comprise a fundamental wave with thefundamental frequency ω₁=1/T₁, which will be indicated as A₁(ω₁), asillustrated in FIG. 3B. The Fourier series further comprises oddharmonic waves A₃(ω₃), A₅(ω₅), . . . , A_(2n+1)(ω_(2n+1)), wherein n=1,2, 3 . . . . Herein, the frequency component A_(2n+1)(ω_(2n+1)) is the(2n+1)-th harmonic wave with respect to the fundamental wave A₁, havinga frequency ω_(2n+1) being equal to (2n+1) times the fundamentalfrequency ω₁. FIG. 3B also shows a part of the 3rd and 5th harmonicwaves.

[0041] It is noted that, although in general the Fourier series is aninfinite series having an infinite number of frequency components, inmost practical circumstances the output signal A from the localoscillator can be approximated very well by a limited number of Fourierterms, for instance five.

[0042] If a polyphase filter would be used to generate quadraturesignals on the basis of such square wave local oscillator signal A, thepolyphase filter will operate on each of said sine-shaped Fouriercomponents in the way described above. Thus, if the polyphase filterwould have a relatively narrow OSR band pass characteristic, onlyaccommodating the fundamental frequency ω₁ of the local oscillator, thepolyphase filter will only generate two sine-shaped output signalsY₁₃(ω₁) and Y₁₄(ω₁) with a mutual phase difference of 90°. Although itis possible to construct square wave signals on the basis of suchsine-shaped output signals, for instance by using an amplifier withlarge gain so that the signals will clip, this will necessitate furthercircuitry, while further small deviations in the sine-shaped outputsignals may lead to important timing deviations in the constructedsquare wave signal. In order to improve the accuracy in thezero-crossings, more Fourier terms should be taken into account.

[0043] If, on the other hand, the polyphase filter would have arelatively wide OSR pass characteristic, also accommodating the harmonicfrequencies ω₃, ω₅, ω₇, etc, the polyphase filter would not generate the90° output signal in a correct way, as will be explained hereinafterwith reference to FIG. 3C.

[0044] First, it is pointed out that the required 90° phase differencebetween the two output signals Y₁₃ and Y₁₄ relates to the fundamentalfrequency ω₁. Thus, in fact, it is required that the two output signalsY₁₃ and Y₁₄ are identical, yet shifted in time over a time distance T₁/4[mod T₁] with respect to each other.

[0045] Second, it is pointed out that, in order to meet the objectivethat the second output signal Y₁₄ is equal to the first output signalY₁₃ yet shifted over T₁/4 [mod T₁], it will be necessary that, whencreating the second output signal Y₁₄, each frequency componentA_(2n+1)(ω_(2n+1)) of the input signal A is shifted over a time distanceT₁/4 [mod T₁]. However, as mentioned above, the polyphase filter 10 hasthe intrinsic property of shifting all frequency componentsA_(2n+1)(ω_(2n+1)) within its pass band over a phase angle of 90°, thisphase shift of 90° being always measured with respect to thecorresponding frequency ω_(2n+1) of the signal component in question.Such a phase shift does not correspond to the required time shift forall frequency components.

[0046] For instance, for the fundamental wave A₁ and the fifth harmonicA₅, the ninth harmonic A₉, etc., a time shift of T₁/4 correspondsrespectively to a phase shift of 90°, 450°, 810°, etc., which is allequivalent to a respective phase shift of +90° [mod 360°]; in otherwords: these are “matching” shifts.

[0047] However, for the third [seventh] { eleventh } harmonic wave A₃[A₇] {A₁₁}, etc., the required time shift of T₁/4 corresponds to arequired phase shift of 270° [630°] {990°}, respectively, in each casebeing equivalent to a required phase shift of −90° [mod 360°]. Asmentioned earlier, the conventional polyphase filter can not deliversuch phase shift. In fact, if a conventional polyphase filter would havea relatively wide OSR pass characteristic, also accommodating theharmonic frequencies ω₃, ω₇, ω₁₁, etc, these harmonic waves are likewiseshifted over +90°, i.e. 180° wrong.

[0048] According to an important aspect of the present invention, thisproblem is overcome if an additional phase shift of 180° is exerted onthe third [seventh] {eleventh} harmonic wave A₃ [A₇] { A₁₁}, etc. Thiscan, for instance, be done by individually selecting the separateharmonic waves, for instance by means of appropriate band pass filters,then individually processing each harmonic wave such that for eachindividual harmonic wave a +90° or a −90° [mod 360°] shift, as required,is obtained, and then combining the shifted harmonic waves.

[0049]FIG. 4 shows an embodiment where the harmonic waves that need tobe shifted over +90° are combined, and one single broadband polyphasefilter 10A is used to perform the +90° shift for all those harmonicwaves in common, whereas the harmonic waves that need to be shifted over−90° are also combined, and one single broadband polyphase filter 10B isused to perform the −90° shift for all those harmonic waves in common.FIG. 4 shows a circuit 1 having an input 3 for receiving an outputsignal A from a local oscillator 2, and two outputs 8 and 9 forgenerating an in-phase output signal I and a quadrature output signal Q,both being identical to the input signal A. As described above, thelocal oscillator signal A is a square wave signal with 50% duty cycle,having a period T₁. As mentioned above, the local oscillator signal Acan be developed into sine-shaped signal components A₁(ω₁), A₃(ω₃),A₅(ω₅), . . . , A_(2n+1)(ω_(2n+1)), wherein n=0, 1, 2, 3 . . . . Thecircuit 1 of FIG. 4 is designed to process four Fourier components, andis suitable for a situation where the output signal A from the localoscillator can be approximated by four Fourier components. From thefollowing description, it will be clear to a person skilled in the arthow this embodiment is to be supplemented in order to take higher orderFourier components into account.

[0050] The circuit 1 comprises a first Fourier component selectionsection 4, selecting those Fourier components which need to be shiftedover +90° in the quadrature output signal Q. These are the Fouriercomponents A₁(ω₁), A₅(ω₅), A₉(ω₉), . . . , A_(2n+1)(ω_(2n+1)), whereinn=0, 2, 4, 6 . . . . For each of those Fourier components, the firstFourier component selection section 4 comprises a corresponding bandpass filter 60 _(2n+1). In the present embodiment, the first Fouriercomponent selection section 4 is intended to select the fundamental waveA₁(ω₁) and the fifth harmonic A₅(ω₅). Thus, the first Fourier componentselection section 4 comprises a first band pass filter 60 ₁ having apass band 61 ₁ in the frequency range between ω₁-BW₁/2 and ω₁+BW₁/2, anda second band pass filter 60 ₅ having a pass band 61 ₅ in the frequencyrange between ω₅-BW₅/2 and ω₅+BW₅/2.

[0051] The input terminals of these band pass filters are connected tothe input 3, while the output terminals of these band pass filters arecoupled to a first adder 71, the output of which is coupled to the firstinput 11A of a first polyphase filter 10A. Thus, the first input 11A ofthis first polyphase filter 10A receives an input signalX_(11A)=A₁(ω₁)+A₅(ω₅). The second input 12A of this first polyphasefilter 10A receives a zero signal.

[0052] Similarly, the circuit 1 comprises further a second Fouriercomponent selection section 5, selecting those Fourier components whichneed to be shifted over −90° in the quadrature output signal Q. Theseare the Fourier components A₃(ω₃), A₇(ω₇), A₁₁(ω₁₁), . . . ,A_(2n+1)(ω_(2n+1)), wherein n=1, 3, 5 . . . . For each of those Fouriercomponents, the second Fourier component selection section 5 comprisescorresponding band pass filters 60 _(2n+1). In the present embodiment,the second Fourier component selection section 5 is intended to selectthe third harmonic A₃(ω₃) and the seventh harmonic A₇(ω₇). Thus, thesecond Fourier component selection section 5 comprises a third band passfilter 60 ₃ having a pass band 61 ₃ in the frequency range betweenω₃-BW₃/2 and ω₃+BW₃/2, and a fourth band pass filter 60 ₇ having a passband 61 ₇ in the frequency range between ω₇-BW₇/2 and ω₇+BW₇/2.

[0053] The input terminals of these band pass filters are connected tothe input 3, while the output terminals of these band pass filters arecoupled to a second adder 72, the output of which is coupled to thesecond input 12B of a second polyphase filter 10B. Thus, the secondinput 12B of this second polyphase filter 10B receives an input signalX_(12B)=A₃(ω₃)+A₇(ω₇). The first input 11B of this second polyphasefilter 10B receives a zero signal.

[0054] The two polyphase filters 10A and 10B are broadband polyphasefilters having, at least for realistic frequencies, (normalized)transfer characteristics H(ω) in accordance with the following equation(3):

H(ω)=1 for ω≧0 and H(ω)=0 for ω<0  (3)

[0055] In fact, the two polyphase filters 10A and 10B may be identical.

[0056] As will be clear from the above explanation of the operation ofthe polyphase filters, the first polyphase filter 10A provides at itsfirst output 13A a first output signal Y_(13A) according to2Y_(13A)=X_(11A)=A₁(ω₁)+A₅(ω₅), and at its second output 14A a secondoutput signal Y_(14A) according to Y_(14A)=jY_(13A). It may be that thefirst output signal Y_(A) has a certain time delay ΔT_(A) with respectto the input signal X_(11A).

[0057] Further, as will also be clear from the above explanation of theoperation of the polyphase filters, the second polyphase filter 10Bprovides at its second output 14B a third output signal Y_(14B)according to 2Y_(14B)=X_(12B)=A₃(ω₃)+A₇(ω₇), and at its first output 13Ba fourth output signal Y_(13B) according to Y_(13B)=−jY_(14B)=jX_(12B).It may be that the third output signal Y_(14B) has a certain time delayΔT_(B) with respect to the input signal X_(12B); the two polyphasefilters 10A and 10B should be matched such that said two time delaysΔT_(A) and ΔT_(B) are equal.

[0058] The first output 13A of the first polyphase filter 10A and thesecond output 14B of the second polyphase filter 10B are coupled to athird adder 73, the output of which is coupled to the first outputterminal 8 of the circuit 1 to provide the first output signal

I=Y _(13A) +Y _(14B)=(A ₁(ω₁)+A ₃(ω₃)+A ₅(ω₅)+A ₇(ω₇))/2

[0059] Similarly, the second output 14A of the first polyphase filter10A and the first output 13B of the second polyphase filter 10B arecoupled to a fourth adder 74, the output of which is coupled to thesecond output terminai 9 of the circuit 1 to provide the second outputsignal

Q=Y _(14A) +Y _(13B) =j(A ₁(ω₁)−A ₃(ω₃)+A ₅(ω₅)−A ₇(ω₇))/2

[0060] The circuit proposed in FIG. 4 functions satisfactorily. However,it needs two polyphase filters. Preferably, the step of shifting thesubsequent harmonic waves over alternatively +90° and −90° is performedby one single polyphase filter that receives all harmonic waves at oneinput.

[0061] According to the present invention, such a design is possiblebecause an additional phase shift of 180° is equivalent to using aFourier component with negative frequency −ω₃, −ω₇, −ω₁₁, etc. in thepolyphase filter, as will be explained with reference to FIGS. 5A-5C.

[0062] Assume that, for a certain positive frequency ω_(x), the transfercharacteristic H(ω) of the polyphase filter 10 obeys the followingequation (4):

H(ω_(x))=0 and H(−ω_(x))=1  (4)

[0063] In FIG. 5A, a first input signal X_(11A)=X(ω_(x)) is applied tothe first input 11, and a second input signal X_(12A)=jX_(11A)=jX(ω_(x))is applied to the second input 12. It follows from the above equations(1) and (4) that the polyphase filter 10 then generates zero outputsignals Y₁₃(ω_(x))=0 and Y₁₄(ω_(x))=0 at its two outputs 13 and 14,respectively.

[0064] In FIG. 5B, a first input signal X_(11B)=X(ω_(x)) is applied tothe first input 11 and a second input signal X_(12B)=jX_(11B)=−jX(ω_(x))is applied to the second input 12. It follows from the above equations(1) and (4) that the polyphase filter 10 then generates a first outputsignal Y_(13B)=Y(ω_(x)) and a second output signalY_(14B)=−jY_(13B)=jY(ω_(x)) at its two outputs 13 and 14, respectively.

[0065] In FIG. 5C, the two input signals used in FIG. 5A and in FIG. 5B,respectively, are added. In other words, the first input 11 receivesX_(11C)=X_(11A)+X_(11B)=2X(ω_(x)), while the second input 12 receivesX_(12C)=X_(12A)+X_(12B)=jX(ω_(x))+(−jX(ω_(x)))=0. Then, also the twooutput signals of FIG. 5A and FIG. 5B are added; therefore, thepolyphase filter 10 then generates a first output signalY_(13C)=jY(ω_(x)) and a second output signal Y_(14C)=−jY_(13C)=jY(ω_(x))at its two outputs 13 and 14, respectively.

[0066] In other words, for such frequencies ω_(x) for which equation (4)applies, if the first input 11 receives a real input signal X(ω_(x))while the second input 12 is zero, then the polyphase filter 10generates first and second output signals Y₁₃(ω_(x))=½·X(ω_(x)) andY₁₄(ω_(x))=−½·jX(ω_(x))=jY₁₃(ω_(x)). These are two real output signalswith −90° phase difference, also indicated as a sine-shaped two-phaseoutput signal having negative frequency.

[0067] Based on this insight, the present invention proposes a polyphasefilter 10 with a novel frequency characteristic 40 as shown in FIG. 6,adapted for use with a local oscillator having a central oscillatorfrequency ω_(LO) and a bandwidth BW_(LO). This frequency characteristic40 has a first OSR band pass region 41 with a positive central frequencyω₁=ω_(LO) and a bandwidth BW₁ substantially equal to the bandwidthBW_(LO) of the local oscillator. More specifically, it is indicated inFIG. 6 that H(ω)=0 in the rejection range 51 between −ω₁−BW₁/2 and−ω₁+BW₁/2, i.e. frequencies around −ω₁ are effectively suppressed. Thefrequency characteristic 40 further has a second OSR band pass region 42with a negative central frequency ω₄₂=−3ω₁ and a bandwidth BW₄₂substantially equal to 3 times the bandwidth BW₁ of the first OSR bandpass region 41. More specifically, it is indicated in FIG. 6 that H(ω)=0in the rejection range 52 between 3ω₁−3BW₁/2 and 3ω₁+3BW₁/2, i.e.frequencies around +3ω₁ are effectively suppressed. In view of thissecond OSR band pass region 42 (and the corresponding rejection region52), the polyphase filter 10 can correctly process the third harmonicwave A₃(ω₃).

[0068] Preferably, the polyphase filter 10 is also designed forcorrectly processing the fifth harmonic wave A₅(ω₅). To that end, thefrequency characteristic 40 further has a third OSR band pass region 43with a positive central frequency ω₄₃=+5ω₁ and a bandwidth BW₄₃substantially equal to 5 times the bandwidth BW₁ of the first OSR bandpass region 41. More specifically, it is indicated in FIG. 6 that H(ω)=0in the rejection range 53 between −5ω₁−5BW₁/2 and −5ω₁+5BW₁/2, i.e.frequencies around −5ω₁ are effectively suppressed. Preferably, thepolyphase filter 10 is also designed for correctly processing theseventh harmonic wave A₇(ω₇). To that end, the frequency characteristic40 further has a fourth OSR band pass region 44 with a negative centralfrequency ω₄₄=−7ω₁ and a bandwidth BW₄₄ substantially equal to 7 timesthe bandwidth BW₁ of the first OSR band pass region 41. Morespecifically, it is indicated in FIG. 6 that H(ω)=0 in the rejectionrange 54 between +7ω₁−7BW₁/2 and +7ω₁+7BW₁/2, i.e. frequencies around+7ω₁ are effectively suppressed.

[0069] In general terms, the polyphase filter 10 has N OSR band passregions, each having a central frequency according toω_(n)=(−1)^((n+1))(2n−1)·ω₁ for n=1, 2, 3, 4, . . . N, and a bandwidthBW_(n) substantially equal to (2n−1) times the bandwidth BW₁ of thefirst OSR band pass region 41.

[0070] In practice, it may be sufficient if N=2, although preferably Nis at least equal to 3. More preferably, N≧5.

[0071] It is observed that, for optimal operational quality, N should beas large as possible. However, in view of the fact that the bandwidth ofthe successive band pass regions increases, N can not be choseninfinitively high; the limit of the possibilities is reached ifneighboring band pass and rejection regions touch, which will be thecase if N=ω₁/BW₁+1.

[0072]FIG. 7 shows a relatively simple embodiment of a polyphase filter110 having the frequency characteristic 40 as described above, whereinN=2, i.e. the polyphase filter 110 of FIG. 7 operates on a fundamentalwave and a third harmonic wave. The polyphase filter 110 has four inputterminals 111, 112, 113, 114, and four corresponding output terminals121, 122, 123, 124. The first (second) [third] {fourth} output terminal121 (122) [123] { 124} is connected to the first (second) [third] {fourth } input terminal 111 (112) [113] { 114} through a first (second)[third] {fourth} transfer channel comprising a series connection of afirst resistor R1 ₁ (R1 ₂) [R1 ₃] {R1 ₄} and a second resistor R2 ₁ (R2₂) [R2 ₃] {R2 ₄}, all first resistors R1 _(i) being substantially equalto each other, and all second resistors R2 _(i) being substantiallyequal to each other.

[0073] The first (second) [third] { fourth } node between the firstresistor R1 ₁ (R1 ₂) [R1 ₃] {R1 ₄} and the second resistor R2 ₁ (R2 ₂)[R2 ₃] {R2 ₄} of the first (second) [third] {fourth} transfer channel isindicated as N1 (N2) [N3] {N4}.

[0074] The first (second) [third] {fourth} input terminal 111 (112)[113] { 114} is coupled to the second (third) [fourth] { first } node N2(N3) [N4] { Ni1} through a first (second) [third] {fourth} first stagecapacitor C12 (C23) [C34] {C41 }, while the second (third) [fourth]{first} node N2 (N3) [N4] {N1} is coupled to the first (second) [third]{fourth } output terminal 121(122) [123] {124} through a first (second)[third] {fourth} second stage capacitor C21 (C32) [C43] {C14}.

[0075] The first and third input terminals 111 and 113 define a firstsignal input. In FIG. 7, an input signal A from a local oscillator 102is received at the first signal input 111, 113 in a balanced way.

[0076] The second and fourth input terminals 112 and 114 define a secondsignal input. In FIG. 7, these terminals are not connected to any signalsource or voltage source, i.e. they are floating. Alternatively, theymay be connected to zero.

[0077] The first and third output terminals 121 and 123 define a firstsignal output. In FIG. 7, the in-phase output signal I is taken fromthese two output terminals 121 and 123. The second and fourth outputterminals 122 and 124 define a second signal output. In FIG. 7, thequadrature output signal Q is taken from these two output terminals 122and 124.

[0078] The polyphase filter 110 of FIG. 7 may be designed for afundamental wave of ω₁ equals about 910 MHz, as used in GSM, forinstance, with a narrow bandwidth of about 40 MHz, while ω₃ is about2730 MHz, by selecting the parameter values approximately as follow:

[0079] R1 ₁ (R1 ₂) [R1 ₃] {R1 ₄}=22 Ω

[0080] R2 ₁ (R2 ₂) [R2 ₃] {R2 ₄}=210.26 Ω

[0081] C12 (C23) [C34] {C41}=2.568 pF

[0082] C21 (C32) [C43] {C14}=820 pF

[0083] It will be clear to a person skilled in the art of designingpolyphase filters how these values are to be amended for obtainingdifferent values for ω₁ and ω₃. Further, it will be clear to a personskilled in the art of designing polyphase filters how the circuit ofFIG. 7 is to be expanded for processing a fifth harmonic wave, a seventhharmonic wave, etc.

[0084] Thus, the present invention succeeds in providing a method anddevice for generating two output signals I; Q each substantiallyidentical to a square-wave input signal A from a local oscillator 2,wherein the first output signal I may have a certain time shift withrespect to the input signal A, and wherein the second output signal Q isshifted over T₁/4 [mod T₁] with respect to the first output signal I, T₁being the period of the input signal A. To generate the first outputsignal I, Fourier components S₁(ω₁), S₃(ω₃), S₅(ω₅), S₇(ω₇), S₉(ω₉),S₁₁(ω₁₁) etc of the input signal A are combined.

[0085] To generate the second output signal Q, Fourier componentsS₁(ω₁), S₅(ω₅), S₉(ω₉) etc of the input signal A are phase shifted over+90° while Fourier components S₃(ω₃), S₇(ω₇), S₁₁(ω₁₁) etc of the inputsignal A are phase shifted over −90°, and the thus shifted Fouriercomponents of the input signal A are combined.

[0086] It should be clear to a person skilled in the art that the scopeof the present invention is not limited to the examples discussed in theabove, but that several amendments and modifications are possiblewithout departing from the scope of the invention as defined in theappending claims. For instance, in the embodiment of FIG. 4, the secondpolyphase filter 10B having a broadband normalized transfercharacteristic

[0087] H(ω)=1 for ω≧0 and H(ω)=0 for ω<0

[0088] can be replaced by a polyphase filter having a broadbandnormalized transfer characteristic H(ω)=1 for ω≦0 and H(ω)=0 for ω>0, inwhich case the output of the second adder 72 will be fed to the firstinput 11B, the second input 12B will receive a zero signal, the firstoutput 13B will be coupled to the first combiner 73, and the secondoutput 14B will be coupled to the second combiner 74.

[0089] Further, it is also possible that, in order to generate thesecond output signal Q, Fourier components S₁(ω₁), S₅((ω₅), S₉(ω₉) etcof the input signal A are phase shifted over −90° while Fouriercomponents S₃(ω₃), S₇(ω₇), S₁₁(ω₁₁) etc of the input signal A are phaseshifted over +90°, and the thus shifted Fourier components of the inputsignal A are combined.

[0090] Further, it is observed that in the above explanation thecharacteristic 40 of the polyphase filter is described only in relationto the passbands or rejection bands around ±ω₁, ±ω₃, ±ω₅, ±ω₇, etc,these bands having a bandwidth of BW₁, 3BW₁, 5BW₁, 7BW₁, respectively,BW₁ being the expected bandwidth of the local oscillator. Forfrequencies between said bands, the characteristic of the polyphasefilter has not been defined. It is noted that, in principle, thecharacteristic of the polyphase filter outside said bands is notcritical. After all, no frequency components are expected in thefrequency regions between said bands. Thus, said bands may in fact bewider than mentioned; the indicated bandwidths are to be considered asminimum widths. Further, although in FIG. 6 the characteristic 40 isillustrated as a combination of pass bands 41, 42, 43, 44 with zerotransfer function in between, the desired functioning may also beobtained by a characteristic that can be described as a broadband passcharacteristic having a certain number of reject regions 51, 52, 53, 54.

[0091] In the claims, any reference signs placed between parenthesesshall not be construed as limiting the claim. The word “comprising” doesnot exclude the presence of elements or steps other than those listed ina claim. The word “a” or “an” preceding an element does not exclude thepresence of a plurality of such elements. The invention can beimplemented by means of hardware comprising several distinct elements,and by means of a suitably programmed computer. In the device claimenumerating several means, several of these means can be embodied by oneand the same item of hardware. The mere fact that certain measures arerecited in mutually different dependent claims does not indicate that acombination of these measures cannot be used to advantage.

1. Method of generating first and second output signals (I, Q)corresponding to an input signal (A) having a fundamental frequency(tol), said output signals (I, Q) having a 90° phase difference withrespect to each other, the method comprising: providing (60 ₁, 60 ₅) apredetermined number of first Fourier components A_(2n+1)(ω_(2n+1)),n=0, 2, 4, . . . , of the input signal (A), wherein ω_(2n+1)=(2n+1)·ω₁;exerting (10A) a 90° phase shift in a first direction (+90°) on saidfirst Fourier components; providing (60 ₃, 60 ₇) a predetermined numberof second Fourier components A_(2n+1)(ω_(2n+1)), n=1, 3, 5, . . . of theinput signal (A); exerting (10B) a 90° phase shift in the oppositedirection (−90°) on said second Fourier components; combining (73) saidpredetermined number of first Fourier components and said predeterminednumber of second Fourier components to provide said first output signal(I); and combining (74) the thus shifted first and second Fouriercomponents to provide said second output signal (Q).
 2. Method accordingto claim 1, wherein said first Fourier components are fed into a circuit(10A) that is adapted to provide an in-line output signal (Y_(13A)) anda +90° shifted output signal (Y_(14A)); wherein said second Fouriercomponents are fed into a circuit (10B) that is adapted to provide anin-line output signal (Y_(14B)) and a −90° shifted output signal(Y_(13B)); wherein said in-line output signals (Y_(13A); Y_(14B)) areadded (73) to provide said first output signal (I); and wherein said+90° shifted output signal (Y_(14A)) and said −90° shifted output signal(Y_(13B)) are added (74) to provide said second output signal (Q). 3.Method according to claim 1, wherein said first Fourier components andsaid second Fourier components are provided by passing the input signal(A) through corresponding band pass filters (60).
 4. Method according toclaim 1, further comprising: providing a polyphase filter (10) having atransfer characteristic H(ω)=1 and H(−ω)=0 for each frequency (ω) incorresponding OSR band pass regions (41, 42, 43, . . . ) around(−1)^(n)·ω_(2n+1) for n=0, 1, 2, 3, . . . , each band pass region (41,42, 43, . . . ) having a bandwidth (BW_(2n+1)) at least equal to (2n+1)times the width (BW₁) of a portion of the first band pass region (41);feeding the input signal (A) to a first input (11) of said polyphasefilter (10); feedin a zero input signal to a second input (12) of saidpolyphase filter (10); taking the first output signal (I) from a firstoutput (13) of said polyphase filter (10); and taking the second outputsignal (Q) from a second output (14) of said polyphase filter (10). 5.Phase-shifting network (1) for generating two output signals (I, Q)corresponding to an input signal (A) at an input (3) of the network, theoutput signals being mutually phase-shifted over 90°, the network (1)comprising: first Fourier component selection means (4) coupled to theinput (3) for selecting at least the fundamental wave A₁(ω₁) from theinput signal (A); second Fourier component selection means (5) coupledto the input (3) for selecting at least the third harmonic wave A₃(ω₃)from the input signal (A); first means (10A) having an input (11A), afirst output (13A), and a second output (14A), the input (11A) beingcoupled to an output of the first Fourier component selection means (4),said first means (10A) being adapted for providing at its first output(13A) a first output signal (Y_(13A)) comprising the Fourier componentsreceived at its input (11A) and for providing at its second output (14A)a second output signal (Y_(14A)) containing the same Fourier componentsas the first output signal (Y_(13A)) but shifted over +90°; second means(10B) having an input (12B), a first output (13B), and a second output(14B), the input (12B) being coupled to an output of the second Fouriercomponent selection means (5), said second means (10B) being adapted forproviding at its second output (14B) a third output signal (Y_(14B))comprising the Fourier components received at its input (12B) and forproviding at its first output (13B) a fourth output signal (Y_(13B))containing the same Fourier components as the third output signal(Y_(14B)) but shifted over −90°; first combiner means (73) having twoinputs coupled with the first output (13A) of said first means (10A) andwith the second output (14B) of said second means (10B), respectively,and having an output coupled with a first output (8) of the network (1);and second combiner means (74) having two inputs coupled with the secondoutput (14A) of said first means (10A) and with the first output (13B)of said second means (10B), respectively, and having an output coupledwith a second output (9) of the network (1).
 6. Phase-shifting network(1) according to claim 5, wherein the first Fourier component selectionmeans (4) comprises a first band-pass filter (60 ₁) having a centralfrequency (ii) and a bandwidth (BW₁), and having an input coupled tosaid input (3); and the second Fourier component selection means (5)comprises a second bandpass filter (60 ₃) having a central frequency(ω₃) substantially equal to three times the central frequency (ω₁) ofthe first band-pass filter (60 ₁), and having a bandwidth (BW₃)substantially equal to three times the bandwidth (BW₁) of the firstband-pass filter (60 ₁), said second band-pass filter (60 ₃) having aninput coupled to said input (3).
 7. Phase-shifting network (1) accordingto claim 6, wherein the first Fourier component selection means (4)further comprises: a third band-pass filter (60 ₅) having a centralfrequency (ω₅) substantially equal to five times the central frequency(ω₁) of the first band-pass filter (60 ₁), and having a bandwidth (BW₅)substantially equal to five times the bandwidth (BW₁) of the firstband-pass filter (60 ₁), said third band-pass filter (60 ₅) having aninput coupled to said input (3); and a first adder (71) having inputscoupled to outputs of the first and third bandpass filters (60 ₁; 60 ₅).8. Phase-shifting network (1) according to claim 7, wherein the secondFourier component selection means (5) further comprises: a fourthband-pass filter (60 ₇) having a central frequency (ω₇) substantiallyequal to seven times the central frequency (ω₁) of the first band-passfilter (60 ₁), and having a bandwidth (BW₇) substantially equal to seventimes the bandwidth (BW₁) of the first bandpass filter (60 ₁), saidfourth band-pass filter (60 ₇) having an input coupled to said input(3); and a second adder (72) having inputs coupled to outputs of thesecond and fourth band-pass filters (60 ₃; 60 ₇).
 9. Phase-shiftingnetwork (1) according to claim 8, wherein the first Fourier componentselection means (4) further comprises: for n=2, 4, 6, . . . : furtherband-pass filters (60 _(2n+1)) having a central frequency (ω_(2n+1))substantially equal to (2n+1) times the central frequency (ω₁) of thefirst band-pass filter (60 ₁), and having a bandwidth (BW_(2n+1))substantially equal to (2n+1) times the bandwidth (BW₁) of the firstband-pass filter (60 ₁), each of said further band-pass filters (60_(2n+1)) having an input coupled to the said input (3) and an outputcoupled to an input of said first adder (71); and wherein the secondFourier component selection means (5) further comprises: for n=3, 5, 7,. . . : further band-pass filters (60 _(2n+1)) having a centralfrequency (ω_(2n+1)) substantially equal to (2n+1) times the centralfrequency (ω₁) of the first band-pass filter (60 ₁), and having abandwidth (BW_(2n+1)) substantially equal to (2n+1) times the bandwidth(BW₁) of the first band-pass filter (60 ₁), each of said furtherband-pass filters (60 _(2n+1)) having an input coupled to the said input(3) and an output coupled to an input of said second adder (72). 10.Phase-shifting network (1) according to claim 5, wherein said firstmeans (10A) comprises a polyphase filter having a broadband normalizedtransfer characteristic H(ω)=1 and H(−ω)=0, and having another input(12A) receiving a zero input signal; and wherein said second means (10B)comprises a polyphase filter having a broadband normalized transfercharacteristic H(ω)=1 and H(−ω)=0, and having another input (11B)receiving a zero input signal.
 11. Phase-shifting network (1; 101),comprising an input (3; 111, 113) for receiving an input signal (A) froma local oscillator (2; 102), the network being adapted for generating,at two outputs (8, 9; 108, 109) two output signals (I, Q) correspondingto the input signal (A) and being mutually phase-shifted over 90°, thenetwork (1; 101) comprising: a polyphase filter (10; 110) having a firstinput (11; 111, 113), a second input (12; 112, 114), a first output (13;121, 123) and a second output (14; 122,124); the first input (11; 111,113) of the polyphase filter (10; 101) being coupled to said input (3),and the second input (12; 112, 114) of the polyphase filter (10; 110)being coupled to receive a zero signal; the first output (13; 121, 123)of the polyphase filter (10; 110) being coupled to said first output (8)for providing the first output signal (I); the second output (14; 122,124) of the polyphase filter (10; 110) being coupled to said secondoutput (9) for providing the second output signal (Q) equal to the firstoutput signal (I) but shifted over 90°; wherein the polyphase filter(10; 110) has a transfer characteristic (40) with a first OSR band-passregion (41) having a central frequency (o)l) and a bandwidth (BW₁) andan associated rejection region (51) having a central frequency (−ω₁) andhaving a bandwidth (BW₅,); and wherein the transfer characteristic (40)comprises a second OSR band-pass region (42) having a central frequencyω₄₂ substantially equal to −3ω₁ and having a bandwidth (BW₄₂), and anassociated rejection region (52) having a central frequencysubstantially equal to 3ω₁ and having a bandwidth (BW₅₂). 12.Phase-shifting network (1) according to claim 11, wherein the transfercharacteristic (40) of said polyphase filter (10) comprises apredetermined number of further OSR band-pass regions, each of suchfurther OSR band-pass regions having a central frequency substantiallyequal to (−1)^(n)·(2n+1)ω₁ and a bandwidth BW_(2n+1), and associatedrejection regions having a central frequency substantially equal to(−1)^(n+1)·(2n+1)ω₁ and a bandwidth BW_(2n+1).
 13. Phase-shiftingnetwork (1) according to claim 11, wherein the central frequency (ω₁) ofthe first OSR band-pass region (41) is substantially equal to thecentral frequency (ω_(LO)) of an expected transmission band of the localoscillator (2; 102), and wherein the bandwidth (BW₁) of the first OSRband-pass region (41) is at least equal to the expected bandwidth(BW_(LO)) of said expected transmission band of the local oscillator (2;102); and wherein the bandwidth BW_(2n+1) of the further OSR passbandregions and corresponding rejection regions is at least equal to (2n+1)times said expected bandwidth (BW_(LO)) of said expected transmissionband of the local oscillator (2; 102).
 14. Device for providing twosquare-wave signals (I; Q) with mutual 90° phase difference, comprisinga network according to claim 5, and a local oscillator (2; 102) coupledto the input (3) of the network (1).
 15. Device according to claim 14,wherein the local oscillator (2; 102) and the network (1) together areimplemented as an integrated circuit in one chip.
 16. Device accordingto claim 14, wherein (ol is substantially equal to the central frequency(ω_(LO)) of the local oscillator (2; 102), and wherein BW₁ issubstantially equal to the bandwidth (BW_(LO)) of the local oscillator(2).
 17. Apparatus for receiving and processing a modulated carrierwave, such as for instance a TV tuner or a telecommunication device, theapparatus comprising a phase-shifting network according to claim 5, or adevice according to claims 14.